Low-noise switching power supply

ABSTRACT

A low-noise, high-power-factor power supply includes a transformer having a primary winding connected to a pair of a.c. input terminals via a rectifier circuit, and a secondary winding connected to a pair of d.c. output terminals via a rectifying and smoothing circuit. Connected between the pair of outputs of the rectifier circuit via at least part of the transformer primary, a switch is turned on and off to keep the d.c. output voltage constant. A smoothing capacitor is connected between the pair of outputs of the rectifier circuit via at least part of the transformer primary and a serial connection of a reverse blocking diode and an inductor. For noise reduction a bypass capacitor is connected between the pair of outputs of the rectifier circuit and in parallel with the serial circuit of the inductor and the reverse blocking diode and at least part of the transformer primary and the smoothing capacitor. The bypass capacitor is less in capacitance than the smoothing capacitor.

BACKGROUND OF THE INVENTION

This invention relates to electric power supplies, and particularly to a switching power supply featuring provisions for attainment of high power factor with a minimum of noise production.

A conversion from an alternating to a direct current is possible by a rectifying and smoothing circuit comprising a rectifying circuit including a diode and connected to an a.c. power supply, and a smoothing circuit including a capacitor and connected to the rectifying circuit. The rectifying and smoothing circuit possesses the disadvantage, however, of being low in power factor as a result of the fact that the smoothing capacitor is charged only at and adjacent the peaks of the a.c. voltage of sinusoidal waveform. Another drawback is that it is incapable of adjustably varying the d.c. output voltage.

Japanese Unexamined Patent Publication No. 8-154379 represents an improvement of the rectifying and smoothing circuit above. It teaches a switching power supply comprising a rectifying circuit, a smoothing capacitor, a d.c.-to-d.c. converter circuit, and an inductive reactor for a higher power factor. The reactor is electrically connected between the pair of output terminals of the rectifying circuit upon closure of a switch included in the d.c.-to-d.c. converter circuit. The desired improvement in power factor is attained as the current flowing through the reactor varies in amplitude in step with the a.c. input voltage.

This prior art switching power supply has proved to have its own shortcomings. Each time the switch of the d.c.-to-d.c. converter circuit opens, the inductor releases the energy that has been stored thereon, with the result that the current flows through the rectifying circuit for charging the smoothing capacitor. The rectifying circuit includes a diode as aforesaid, to which diode the current due to the energy release from the reactor flows at a repetition rate of as high as 20 to 150 kilohertz. Abrupt changes in the magnitude of the current flowing through the diode are known to give rise to noise with a frequency much higher than that at which the switch is turned on and off. A noise filter, sometimes referred to as line filter, has conventionally been connected between the a.c. input terminals and the rectifying circuit in order to prevent the leakage of the high frequency noise produced by the rectifying circuit and by the d.c.-to-d.c. converter circuit.

The trouble has been the high-frequency noise due to the diode of the rectifying circuit. The total resulting noise has been of annoyingly high level, requiring the provision of several noise filters which have added substantively to the size and manufacturing cost of this type of switching power supply.

SUMMARY OF THE INVENTION

The present invention seeks to minimize the noise production of switching power supplies of the kind defined, without in any way adversely affecting their power factor in so doing.

Briefly, the invention may be summarized as a switching power supply capable of translating a.c. voltage into d.c. voltage, comprising a pair of a.c. input terminals for inputting a.c. voltage, a pair of d.c. output terminals for outputting d.c. voltage, a rectifier circuit connected to the pair of input terminals, a transformer having a winding, a rectifying and smoothing circuit connected between the transformer and the pair of d.c. output terminals, an inductor for improvement of the power factor of the input terminals, a reverse blocking diode, a smoothing capacitor connected between the pair of outputs of the rectifier circuit via at least part of the transformer winding, the reverse blocking diode and the inductor, a switch connected between the pair of outputs of the rectifier circuit via at least the inductor and the reverse blocking diode and in parallel with the smoothing capacitor via at least part of the transformer winding, a switch control circuit connected to the switch for on-off control of the switch at a repetition frequency higher than the frequency of the a.c. input voltage, and a bypass capacitor. The bypass capacitor is connected between the pair of outputs of the rectifier circuit and in parallel with the serial circuit of the inductor and the reverse blocking diode and at least part of the transformer winding and the smoothing capacitor. The bypass capacitor is less in capacitance than the smoothing capacitor. The reverse blocking diode has a reverse recovery time shorter than the nonconducting periods of the switch,

The invention particularly features the bypass capacitor, with a capacitance less than that of the smoothing capacitor, which is connected between the pair of outputs of the rectifier circuit. Consequently, unlike the prior art, the current due to energy release from the inductor during the nonconducting periods of the switch does not flow through the rectifier circuit, but through the bypass capacitor taught by the invention. High frequency noise production by the rectifier circuit is avoided as the intermittent current flow through the diode of the rectifier circuit is prevented as above.

The bypass capacitor is so connected, as summarized above, that at least part of the transformer winding is interposed between the inductor and the smoothing capacitor. The smoothing capacitor can then be charged with a relatively low voltage and so need not be of expensive construction for withstanding high voltages.

For further noise reduction an additional capacitor may be connected in parallel with the switch at least via the reverse blocking diode, and with the bypass capacitor via the inductor. This additional capacitor will redound for reduction of high frequency noise due to the operation of the switch.

The above and other objects, features and advantages of this invention will become more apparent, and the invention itself will best be understood, from a study of the following description and appended claims, with reference had to the attached drawings showing the preferred embodiments of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic electrical diagram, partly in block form, of the switching power supply constructed according to the novel concepts of the present invention;

FIG. 2 is a schematic electrical diagram of the noise filter included in the FIG. 1 power supply;

FIG. 3 is a schematic electrical diagram of the switch control circuit included in the FIG. 1 power supply;

FIG. 4, consisting of (A) through (F), is a series of waveforms useful in explaining the operation of the FIG. 1 power supply;

FIG. 5 is a diagram similar to FIG. 1 but showing a second preferred form of switching power supply according to the invention;

FIG. 6 is a diagram similar to FIG. 1 but showing a third preferred form of switching power supply according to the invention;

FIG. 7 is a diagram similar to FIG. 1 but showing a fourth preferred form of switching power supply according to the invention;

FIG. 8 is a diagram similar to FIG. 1 but showing a fifth preferred form of switching power supply according to the invention;

FIG. 9 is a diagram similar to FIG. 1 but showing a sixth preferred form of switching power supply according to the invention;

FIG. 10 is a diagram similar to FIG. 1 but showing a seventh preferred form of switching power supply according to the invention;

FIG. 11 is a diagram similar to FIG. 1 but showing an eighth preferred form of switching power supply according to the invention;

FIG. 12 is a schematic electrical diagram equivalently depicting in more detail the two switches included in the FIG. 11 power supply; and

FIG. 13, consisting of (A) through (G), is a series of waveform diagrams useful in explaining the operation of the FIGS. 11 and 12 power supply.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The switching power supply shown in FIG. 1 by way of a preferable embodiment of the invention has a pair of input terminals 1 and 2 which are to be connected to a source, not shown, of commercial alternating current with a frequency of, for instance, 50 Hz. A noise filter 3 is connected to this pair of input terminals 1 and 2.

As shown in detail in FIG. 2, the noise filter 3 comprises two capacitors C_(a) and C_(b) connected respectively between the input terminals 1 and 2 and the ground, another capacitor C_(c) connected between these input terminals, two inductors L_(a) and L_(b) connected respectively to the input terminals 1 and 2 via the a.c. conductors 1 _(a) and 1 _(b), and still another capacitor C_(d) connected between these a.c. conductors.

Referring back to FIG. 1, the noise filter 3 of the foregoing construction is connected by way of the pair of a.c. conductors 1 _(a) and 1 _(b) to a bridge rectifier circuit 4 having four diodes D₁, D₂, D₃ and D₄ in bridge connection. A junction 4 _(a) between the diodes D₁ and D₂ is connected to the first a.c. input conductor 1 _(a) by way of a first input of this rectifier circuit 4, and a junction 4 _(b) between the diodes D₃ and D₄ to the second a.c. input conductor 1 _(b) by way of a second input. The first output, then, of the rectifier circuit 4 is a junction 4 _(c), between the diodes D₁ and D₃, and the second output thereof a junction 4 _(d) between the diodes D₂ and D₄. All that is required for the diodes D₁-D₄ is to rectify the 50 Hz a.c. voltage, it being unnecessary for them to turn on and off in response to the switching of this power supply.

The pair of outputs 4 _(c), and 4 _(d) of the rectifier circuit 4 are connected to a transformer 5 via an inductor L₁, a bypass capacitor C₁, another capacitor C₂ for elimination of the high frequency component, a reverse blocking diode D₅, and a semiconductor switch Q₁. The transformer 5 has a primary winding N₁, and a secondary winding N₂ which are electromagnetically coupled together, and a magnetic core. The transformer primary N₁, is center tapped at 8 and thereby divided into two parts N_(1a) and N_(1b). The transformer primary N₁, and secondary N₂ are opposite in polarization, as indicated by the dots in FIG. 1.

The smoothing capacitor C_(dc), preferably an electrolytic capacitor, has one of its opposite polarity terminals connected to the first output 4 _(c), of the rectifier circuit 4 via the transformer primary first part N_(1a), reverse blocking diode D₅ and inductor L₁. The other terminal of the smoothing capacitor C_(dc) is connected to the second output 4 _(d) of the rectifier circuit 4. Notwithstanding the solid-line showing of FIG. 1, however, the inductor L₁ and reverse blocking diode D₅ could be connected between the second mentioned terminal of the smoothing capacitor C_(dc) and the second output 4 _(d) of the rectifier circuit 4, as indicated by the broken lines in the same figure.

Shown as an insulated gate field effect transistor, the switch Q₁ is connected in parallel with the smoothing capacitor C_(dc) via the transformer primary N₁. Additionally, the switch Q₁ is connected to the first rectifier output 4 _(c), via the transformer primary second part N_(1b), reverse blocking diode D₅ and inductor L₁ on one hand and, on the other hand, directly to the second rectifier output 4 _(d). The reverse blocking diode D₅ is designed to go on and off in synchronism with the switch Q₁, with a reverse recovery time shorter than each nonconducting period of the switch Q₁.

The transformer secondary N₂ has its opposite extremities connected respectively to the pair of output terminals 10 and 11 via a rectifying and smoothing circuit 6. The rectifying and smoothing circuit 6 comprises a rectifying diode D₀ and a smoothing capacitor C₀. Connected between one extremity of the transformer secondary N₂ and the output terminal 10, the rectifying diode D₀ is so oriented as to be conductive when the switch Q₁ is off, and nonconductive when the switch Q₁ is on. The capacitor C₀ is connected in parallel with the transformer secondary N₂ via the diode D₀. A unidirectional output voltage is thus obtained between the pair of output terminals 10 and 11 for feeding a load, not shown, connected thereto.

The present invention particularly features the bypass capacitor C₁ connected between the outputs 4 _(c), and 4 _(d) of the rectifier circuit 4. The bypass capacitor C₁ is less, preferably not more than one hundredth, in capacitance than the smoothing capacitor C_(dc). With such small capacitance, the bypass capacitor C₁ is practically incapable of smoothing the output from the rectifier circuit 4; instead, the voltage across this capacitor C₁ changes with the output from the rectifier circuit 4.

In order to filter out the high frequency component of the incoming supply current, the capacitor C₂ is connected in parallel with the bypass capacitor C₁ via the inductor L₁ and, additionally, in parallel with the switch Q₁ via the reverse blocking diode D₅ and transformer primary second part N_(1b). Much less in capacitance than the smoothing capacitor C_(dc) or the bypass capacitor C₁, the capacitor C₂ is designed to absorb the high frequency noise that may be caused by the switch Q₁, diodes D₀ and D₅, etc., on the output side of that capacitor C₁.

As shown also in FIG. 1, a switch control circuit 7 has inputs connected to the pair of output terminals 10 and 11 via conductors 12 and 13, respectively, and an output connected to the control terminal of the switch Q₁ via a conductor 14. The switch control circuit 7 is designed to turn the switch Q₁ on and off with a repetition frequency required to keep the voltage between the pair of output terminals 10 and 11 at a required value.

FIG. 3 is a more detailed illustration of the switch control circuit 7. Included is a serial connection of two voltage dividing resistors 15 and 16 connected between the pair of input conductors 12 and 13. The junction between the resistors 15 and 16 is connected to one input of a differential amplifier 18, the other input of which is connected o a reference voltage source 17. The output of the differential amplifier 18 is connected to one input of a comparator 20, the other input of which is connected to a sawtooth generator circuit 19. The output of the comparator 20 is connected to the control terminal of the switch Q₁, FIG. 1, by way of the output conductor 14.

The sawtooth generator 19 puts out a sawtooth voltage with a frequency (e.g. 20 kHz) that is higher than the frequency (e.g. 50 Hz) of the a.c. voltage V_(ac) between the pair of input terminals 1 and 2. Thus the comparator 20 puts out a series of duration modulated switch control pulses in synchronism with the sawtooth voltage, making on-off control of the switch Q₁ accordingly. As required or desired, the differential amplifier 18 and the comparator 20 may be coupled photoelectrically, as by a light emitting diode and phototransistor, instead of directly as in FIG. 3.

Operation

In use the pair of a.c. input terminals 1 and 2 are to be connected to an unshown source of a.c. power, and the pair of d.c. output terminals 10 and 11 to an unshown load. The smoothing capacitor C_(dc) will be charged to the desired d.c. voltage V_(c). The resulting steady state operation of this representative switching power supply will be discussed hereinbelow with reference to FIG. 4 which shows he waveforms appearing in various parts of FIG. 1.

At (A) in FIG. 4 is shown a series of switch control pulses V_(g1) impressed to the control terminal of the switch Q₁ for its on-off control. During each conducting period T_(on) of the switch Q₁, as from t₂ to t₃ in FIG. 4, current will flow through a first path comprising the first rectifier output 4 _(c), inductor L₁, reverse blocking diode D₅, transformer primary second part N_(1b), switch Q₁, and second rectifier output 4 _(d), as well as through a second path comprising the smoothing capacitor C_(dc), transformer primary N₁, and switch Q₁. Consequently, there flows through the conducting switch Q₁ the resultant of the current that flowed through the inductor L₁ and the current that came from the smoothing capacitor C_(dc). This combined current flowing through the switch Q₁ is shown at (C) in FIG. 4 and therein designated I_(q1), Energy is stored on both inductor L₁ and transformer 5 during each such conducting period T_(on) of the switch Q₁.

The switch Q₁ is shown to go off at t₃ in FIG. 4 and remain so until t₅. During each such nonconducting period T_(off) of the switch Q₁, when the switch current I_(q1), is zero as at (C) in FIG. 4, the reverse blocking diode D₅ is conductive as long as its anode potential is higher than the cathode potential. A current flows therefore through the path comprising the first rectifier output 4 _(c), inductor L₁, reverse blocking diode D₅, transformer primary first part N_(1a), smoothing capacitor C_(dc), and second rectifier output 4 _(d), charging the capacitor C_(dc). Also, during each such nonconducting period T_(off), the energy that has been stored on the inductor L₁ will be released, with the result that the current I₁ flowing through this inductor dwindles with time, as at (B) in FIG. 4, until it becomes zero at t₄ which is shortly before t₅. The transformer 5 also releases its energy storage during the nonconducting period T_(off), causing a voltage to be developed across the transformer secondary N₂ with the consequent conduction of the diode D₀. The conduction of the diode D₀ results in turn in current flow through the path comprising the transformer secondary N₂, diode D₀, capacitor C₀, and load. The voltage across the capacitor C₀ clamps the transformer secondary N₂ during the conduction of the diode D₀, so that a voltage is induced across the transformer primary N₁, with a magnitude depending upon the ratio of the turns of the transformer windings. This voltage is of the same orientation as the voltage V_(c) across the smoothing capacitor C_(dc). The voltage V_(c) is defined as:

V _(c) =V ₄ +V _(L1) −V _(N1a)

where

V₄=the output voltage of the rectifier circuit 4,

V_(L1)=the voltage across the inductor L₁,

V_(N1a)=the voltage across the first part N_(1a) of the transformer primary N₁.

The smoothing capacitor C_(dc) is charged to a voltage as low as less than the sum of the rectifier output voltage V₄ and the inductor voltage V_(L1). The current ceases to flow through the reverse blocking diode D₅ at t₄ in FIG. 4 when the inductor L₁ completes its energy release. However, the current I₄ from the rectifier circuit 4 is not cut off at this moment by reason of the connection of the bypass capacitor C₁ and the high frequency component elimination capacitor C₂ between the pair of rectifier outputs 4 _(c) and 4 _(d). The rectifier output current I₄ continues to flow into these capacitors C₁ and C₂ during the ensuing t₄-t₅ period.

The rectifier output current I₄ is depicted idealized (i.e. without ripple) at (D) in FIG. 4. Both this rectifier output current I₄ and the a.c. input current I_(ac), at (E) in FIG. 4, flow during the t₁-t₆ period of the first 180 electrical degrees, from t₀ to t₇, of the a.c. input voltage V_(ac), at (F) in FIG. 4 and during the t₈-t₉ period of the second 180 electrical degrees, from t₇ to t₁₀, of the a.c. input voltage V_(ac). Both inductor current I₁, and rectifier output current I₄ vary in amplitude with the a.c. input voltage V_(ac) and flow during the relatively long periods of from t₁ to t₆ and from t₈ to t₉, resulting in close approximation of the current waveform I_(ac) at the a.c. input terminals 1 and 2 to a sinusoidal wave and in improvement in power factor.

The output from the differential amplifier 18, FIG. 3, of the switch control circuit 7 will lower when the output voltage of the FIG. 1 power supply rises above the desired value. The output pulses of the comparator 20 will then shorten in duration, only to an extent necessary to return the output voltage to normal. Conversely, upon decrease in the power supply output voltage, the output from the differential amplifier 18 will rise, making the comparator output pulses longer in duration, until the output voltage returns to normal.

The advantages gained by this embodiment of the invention may be recapitulated as follows:

1. The current due to energy release by the inductor L₁, which is provided for improvement of power factor, flows through the circuit comprising the inductor L₁, transformer primary first part N_(1a), smoothing capacitor C_(dc) and bypass capacitor C₁, but not through the rectifier circuit 4. The current of the four diodes D₁-D₄ of the rectifier circuit 4 has a low frequency (e.g. 50 Hz) in synchronism with the a.c. input voltage V_(ac), as indicated at (D) in FIG. 4, so that these diodes do not produce noise having frequencies equal to the switching frequency or even higher. The noise filter 3 can therefore be of no such inconveniently large size as has been required heretofore.

2. The switch Q₁ combines the functions of switching the voltage applied to the transformer primary N₁, for d.c.-to-d.c. conversion and of switching the current flowing to the inductor L₁ for improvement of the power factor.

3. The inductor L₁ is connected to the smoothing capacitor C_(dc) not directly but via the first part N_(1a) of the transformer primary N₁, so that the voltage V_(c) for charging the smoothing capacitor can be lower than the sum of the output voltage V₄ of the rectifier circuit 4 and the voltage V_(L1) across the inductor L₁. The smoothing capacitor C_(dc) need not be of expensive make capable of withstanding high voltages.

Embodiment of FIG. 5

The switching power supply of FIG. 5 differs from that of FIG. 1 in that the transformer primary N₁, is untapped, and that the inductor L₁ is connected via the reverse blocking diode D₅ to the junction between transformer primary N₁ and the drain of the FET switch Q₁. All the other details of construction of the FIG. 5 power supply are as previously set forth with reference to FIGS. 1-3.

Thus the FIG. 5 power supply is so modified that no part of the transformer primary N₁, intervenes between inductor L₁ and switch Q₁ The current I₁ flowing through the inductor L₁ during the conducting periods of the switch Q₁ is of greater magnitude than in the FIG. 1 embodiment, realizing further improvement in power factor. An additional advantage is that a materially less voltage is needed for charging the capacitor C_(dc) as the transformer primary N₁, wholly exists between inductor L₁ and capacitor C_(dc). The noise reduction capability of this modified power supply is the same as that of the FIG. 1 embodiment.

Embodiment of FIG. 6

The device of FIG. 6 differs from that of FIG. 1 only in additionally comprising a capacitor C₃ and a diode D₆. The capacitor C₃ is connected between reverse blocking diode D₅ and the junction between the transformer primary N₁, and the drain of the FET switch Q₁ for use as a bias power supply. The diode D₆ is connected between the capacitor C₃ and the tap 8 on the transformer primary N₁, and is so oriented that the capacitor C₃ will be charged with the polarity indicated in FIG. 6.

Such being the modified construction of the FIG. 6 switching power supply, the capacitor C₃ will be charged due to the voltage developing across the transformer primary N₁ during the nonconducting periods of the switch Q₁, that is, by the current flowing through the path comprising the transformer primary second part N_(1b), capacitor C₃. and diode D₆. During the conducting periods of the switch Q₁, on the other hand, the current I_(q1) will flow through the path comprising the rectifier circuit 4, inductor L₁, reverse blocking diode D₅, biasing capacitor C₃, and switch Q₁ The current I_(q1) will be of relatively great magnitude even when the sinusoidal output voltage from the rectifier circuit 4 is relatively low, because the voltage across the biasing capacitor C₃ is added to the rectifier output voltage V₄. Remarkable improvement can therefore be made in power factor.

Embodiment of FIG. 7

The biasing capacitor C₃ need not be connected to the junction between transformer primary N₁ and switch Q₁ as in FIG. 6 but, as indicated in FIG. 7, to a tap 8 _(a) on the second part N_(1b) of the transformer primary N₁. This FIG. 7 embodiment is akin to that of FIG. 6 in all the other details of construction.

The position of the tap 8 a on the transformer primary second part N_(1b) is variable for adjustment of the voltage across the biasing capacitor C₃, of the magnitude of the current I_(q1) during the conducting periods of the switch Q₁, and of the voltage for charging the smoothing capacitor C_(dc) during the nonconducting periods of the switch Q₁. As an additional modification of this FIG. 7 device, the diode D₆ could be connected between the capacitor C₃ and the junction between the smoothing capacitor C_(dc) and the transformer primary N₁ as indicated by the broken lines in this figure.

Embodiment of FIG. 8

The transformer 5 of the FIG. 1 embodiment may be modified as shown at 5 _(a) in FIG. 8. The modified transformer 5 _(a) differs from its FIG. 1 counterpart 5 in that the first part N_(1a) of the primary winding N₁, of the transformer 5 _(a) is opposite in polarization to that of the primary winding of the transformer 5. In conformity with this change in polarities the smoothing capacitor C_(dc) is connected to the tap 8, with the result that a serial connection of the transformer primary second part N_(1b) and switch Q₁ is connected in parallel with the smoothing capacitor C_(dc). The reverse blocking diode D₅ has its cathode connected to the smoothing capacitor C_(dc) via the transformer primary first part N_(1a). Also, between the pair of outputs 4 _(c), and 4 _(d) of the rectifier circuit 4, there is connected a serial network of the inductor L₁, reverse blocking diode D₅, transformer primary N₁, and switch Q₁.

The desired improvement in power factor is attained as current flows during the conducting periods of the switch Q₁, through the path comprising the rectifier circuit 4, inductor L₁, diode D₅, transformer primary N₁, and switch Q₁, as well as through the path comprising the smoothing capacitor C_(dc), transformer primary second part N_(1b), and switch Q₁. The energy that has been stored on the transformer during each conducting period of the switch Q₁ is released upon opening of the switch, causing conduction through the diode D₀ of the rectifying and smoothing circuit 6. The energy that has been stored on the inductor L₁, on the other hand, is released through the path comprising the inductor L₁, diode D₅, transformer primary first part N_(1a), smoothing capacitor C_(dc), and bypass capacitor C₁, thereby charging the capacitor C_(dc). The voltage across the transformer primary first part N_(1a) is opposite in polarity to both the output voltage V₄ of the rectifier circuit 4 and the voltage V_(L1) across the inductor L₁, the smoothing capacitor C_(dc) is charged to a voltage less than the sum of these voltages V₄ and V_(L1).

Embodiment of FIG. 9

The switching power supply of FIG. 9 incorporates another modified transformer 5 _(b) and a modified rectifying and smoothing circuit 6 _(a) but is akin to the FIG. 1 embodiment in all the other respects. The transformer 5 _(b) has a primary winding N₁, and secondary winding N₂ of the same polarization, as indicated by the dots in this figure, so that a forward d.c.-to-d.c. converter is constituted of the transformer 5 _(b), switch Q₁, and rectifying and smoothing circuit 6 _(a). The rectifying and smoothing circuit 6 _(a) is itself of conventional make comprising the diode D₀, smoothing inductor L₂, smoothing capacitor C₀, and commutation diode D₈. The diode D₀ conducts during the conducting periods of the switch Q₁.

Embodiment of FIG. 10

Still another modified transformer 5 _(c) and another modified rectifying and smoothing circuit 6 _(b) are both incorporated in the device of FIG. 10, which is identical with the FIG. 1 device in all the other respects. The transformer 5 _(c) differs from its FIG. 1 counterpart 5 in that not just the primary winding N₁, but the secondary winding N₂ too is center tapped, as at 21, and so divided into a pair of halves N_(2a) and N_(2b). The modified rectifying and smoothing circuit 6 _(b) comprises two diodes D₀ and D_(0a) and the capacitor C₀. The first half N_(2a) of the transformer secondary N₂ is connected in parallel with the smoothing capacitor C₀ via the diode D₀, and the second half N_(2b) in parallel with the smoothing capacitor C₀ via the other diode D_(0a). As required, an inductor, not shown, may be connected between the diodes D₀ and D_(0a) and the smoothing capacitor C₀.

Embodiment of FIGS. 11-13

The switching power supply of FIG. 11 differs from that of FIG. 1 in additionally comprising a resonance capacitor C_(x), a resonance switch Q₂ in the form of a field effect transistor, and a switch control circuit 30 for the resonance switch Q₂. All the other details of construction are as previously set forth with reference to FIGS. 1-3.

The resonance switch Q₂ is connected in parallel with the power control switch Q₁ via the resonance capacitor C_(x). Both switches Q₁, and Q₂ take the form of insulated gate field effect transistors, each with the source connected to the substrate or bulk. Therefore, as depicted in FIG. 12, the switches Q₁ and Q₂ can be shown as switches proper S₁ and S₂ having diodes D₁₁, and D₁₂ connected inversely in parallel therewith, as well as stray capacitances C₁₁ and C₁₂, connected in parallel therewith. Alternatively, however, the diodes D₁₁, and D₁₂ and capacitors C₁₁ and C₁₂ may be provided as discrete units.

FIG. 13 indicates at (A) and (C) the drain-source voltages V_(q1) and V_(q2) of the switches Q₁ and Q₂, at (B) and (D) the control signals V_(g1) and V_(g2) applied from the switch control circuits 7 an 30 to the gates of the switches Q₁ and Q₂, at (E) and (F) the currents I_(s1), and I_(s2) flowing through the switches proper S₁ and S₂ of the switches Q₁, and Q₂, and at (G) the voltage V_(cx) across the resonance capacitor C_(x).

A comparison of the waveforms (B) and (D) in FIG. 13 will reveal that the resonance switch control signal V_(g2) is designed to open the resonance switch Q₂ during the nonconducting periods of the power control switch Q₁ under the control of the switch control signal V_(gl). It will be further observed from these waveforms that the durations of the switch control pulses V_(g2) are shorter than the spacings between the power control switch control pulses V_(g1), and vice versa. Consequently, between the conducting periods of the switches Q₁ and Q₂, there exist brief transition periods during which both of these switches are off, as from t₂ to t₃ and from t₄ to t₅ in FIG. 13.

During the conducting periods of the switch Q₁, as from t₁ to t₂ in FIG. 13, current flows through the closed circuit comprising the capacitor C_(dc), transformer primary N₁, and switch Q₁, as well as through the closed circuit comprising the capacitor C₁, inductor L₁, diode D₅, transformer primary second part N_(1b), and switch Q₁. When the switch Q₁ is subsequently opened t₂, current will flow through the stray capacitance C₁₁, FIG. 12, of the switch Q₁ thereby charging the same. The voltage V_(q1) across the switch Q₁ will therefore rise gradually during the ensuing t₂-t₃ period, as at (A) in FIG. 13. Thus has been accomplished the zero voltage switching of the switch Q₁, when it is turned off, with the consequent reduction of switching loss.

Upon completion of the charging of the stray capacitance C₁₁, at t₃, current will flow through the closed circuit of the capacitor C_(dc), transformer primary N₁, resonance capacitor C_(x), and the switch S₂ or diode D₁₂ of the resonance switch Q₂, as well as through the closed circuit of the capacitor C₁, inductor L₁, diode D₅, transformer primary second part N_(1b), capacitor C_(x), and switch S₂ or diode D₁₂. Then, as the capacitor C_(x) discharges, current will flow reversely through the closed circuit of the capacitor C_(x), transformer primary N₁, capacitor C_(dc), and switch S₂, as well as through the circuit of the capacitor C₁, inductor L₁, diode D₅, transformer primary first part N_(1a), and capacitor C_(dc). The capacitor C_(x) is of such great capacitance that, after having been charged with its right hand side, as seen in FIGS. 11 and 12, positive, it will maintain this voltage V_(cx) approximately constant as at (G) in FIG. 13.

When the switch Q₂ is turned off at t₄, the stay capacitance C₁₁, of the switch Q₁, will be reversely charged, with the result that the voltage V_(q1) across the switch Q₁, will drop as at (A) in FIG. 13. The current reversely charging the stray capacitance C₁₁, will flow through the closed circuit of the capacitor C_(dc), capacitance C₁₁, and transformer primary N₁, so that the charge on the capacitance C₁₁ will be returned to the capacitor C_(dc), or to the capacitor C₀ of the rectifying and smoothing circuit 6. The voltage across the switch Q₁ will be approximately zero when it is turned on at t₅ to permit the current I_(q1) to flow therethrough, again realizing a decrease in switching loss.

Possible Modifications

Notwithstanding the foregoing detailed disclosure it is not desired that the present invention be limited by the exact details of the illustrated embodiments. The following is a brief list of possible modifications or alterations that will readily suggest themselves to one skilled in the art on the basis of this disclosure and without departure from the scope of the invention:

1. The second switch Q₂ together with the capacitor C_(x) and switch control circuit 30 could be added to all of the FIGS. 5-10 embodiments.

2. The transformers 5 and 5 a and rectifying and smoothing circuit 6 of the FIGS. 5-8 embodiments could all be of the forward constructions like their FIG. 9 counterparts 5 _(b) and 6 _(a).

3. A bipolar transistor and like semiconductor switch other than an FET could be used as the switch Q₁.

4. The rectifier circuit 4 and switch Q₁, could be reversed in polarity.

5. The inductor L₁ and reverse blocking diode D₅ could be connected in positions indicated by the broken lines in FIGS. 1 and 5-11.

6. Autotransformers could be employed in places of the transformers 5, 5 _(a), 5 _(b) and 5 _(c). 

What is claimed is:
 1. A switching power supply capable of translating a.c. voltage into d.c. voltage, comprising: (a) a pair of a.c. input terminals for inputting a.c. voltage having a known frequency; (b) a pair of d.c. output terminals for outputting d.c. voltage; (c) a rectifier circuit connected to the pair of input terminals; (d) a transformer having a winding; (e) a rectifying and smoothing circuit connected between the transformer and the pair of d.c. output terminals for providing the d.c. output voltage; (f) an inductor for improvement of the power factor of the input terminals; (g) a reverse blocking diode; (h) a smoothing capacitor connected between the pair of outputs of the rectifier circuit via at least part of the transformer winding, the reverse blocking diode and the inductor; (i) a switch connected between the pair of outputs of the rectifier circuit via at least the inductor and the reverse blocking diode, and in parallel with the smoothing capacitor via at least part of the transformer winding; (j) a switch control circuit connected to the switch for on-off control of the switch at a repetition frequency higher than the frequency of the a.c. input voltage; (k) a bypass capacitor for reduction of high frequency noise connected between the pair of outputs of the rectifier circuit and in parallel with the serial circuit of the inductor and the reverse blocking diode and at least part of the transformer winding and the smoothing capacitor, the bypass capacitor being less in capacitance than the smoothing capacitor; and (l) the reverse blocking diode having a reverse recovery time shorter than the nonconducting periods of the switch.
 2. The switching power supply of claim 1 further comprising an additional capacitor for further reduction of high frequency noise connected in parallel with the switch via at least the reverse blocking diode and with the bypass capacitor via the inductor, the additional capacitor being less in capacitance than the smoothing capacitor.
 3. The switching power supply of claim 1 wherein the transformer winding is tapped to provide a first and a second division, wherein the first division of the transformer winding forms a serial circuit with the inductor and the reverse blocking diode and the switch, which serial circuit is connected between the pair of outputs of the rectifier circuit, and wherein the smoothing capacitor is connected in parallel with a serial circuit of both first and second divisions of the transformer winding and the switch.
 4. The switching power supply of claim 1 wherein the complete transformer winding forms a serial circuit with the inductor and the reverse blocking diode and the smoothing capacitor, which serial circuit is connected between the pair of outputs of the rectifier circuit, and wherein a serial circuit of the complete transformer winding and the switch is connected in parallel with the smoothing capacitor.
 5. The switching power supply of claim 1 further comprising: (a) an additional capacitor connected in series with the inductor and the reverse blocking diode; and (b) means connected to the transformer winding and the additional capacitor for charging the latter by utilizing a voltage that develops across the transformer winding during the nonconducting periods of the switch.
 6. The switching power supply of claim 1 wherein the transformer winding is tapped to provide a first and a second division which are opposite in polarity, wherein the first division of the transformer winding forms a serial circuit with the inductor and the reverse blocking diode and the smoothing capacitor, which serial circuit is connected between the pair of outputs of the rectifier circuit, and wherein the second division of the transformer winding forms a second serial circuit with the switch, which second serial circuit is connected in parallel with the smoothing capacitor.
 7. The switching power supply of claim 1 wherein the transformer comprises a primary winding and a secondary winding electromagnetically coupled to the primary winding, and wherein the rectifying and smoothing circuit is connected to the secondary winding and adapted to liberate energy from the transformer during the nonconducting periods of the switch.
 8. The switching power supply of claim 1 wherein the transformer comprises a primary winding and a secondary winding electromagnetically coupled to the primary winding, and wherein the rectifying and smoothing circuit is connected to the secondary winding and adapted to liberate energy from the transformer during the conducting periods of the switch.
 9. The switching power supply of claim 1 wherein the transformer comprises a primary winding and a secondary winding electromagnetically coupled to the primary winding, and wherein the secondary winding is connected to the rectifying and smoothing circuit and tapped to provide a first and a second division, and wherein the rectifying and smoothing circuit comprises: (a) a first and a second diode connected respectively to the first and the second division of the transformer secondary; and (b) a smoothing capacitor connected to the first and the second diode.
 10. The switching power supply of claim 1 further comprising: (a) a capacitor connected in parallel with the switch; (b) a first resonant diode inversely connected in parallel with the switch; (c) a resonant capacitor; (d) a second switch connected in parallel with the first recited switch via the resonant capacitor; (e) a second resonant diode connected in parallel with the second switch and being opposite in orientation to the first resonant diode; and (f) a second switch control circuit connected to the second switch for holding the latter closed from a moment following the beginning of each nonconducting period of the first switch to a moment preceding the end of each nonconducting period of the first switch. 